N94- 22746 SAW Based Systems for Mobile Communications Satellites R.C. Peach, N. Miller and M. Lee COM DEV Ltd. 155 Sheldon Drive Cambridge, Ontario N1R 7H6 (519) 622-2300 (519) 622-1691 ABSTRACT (BSSF), or seamless combining, which allows a significant recovery of guard band spectrum [1] [2]. The Modern mobile communications satellites, such as principle of this method is to use banks of contiguous INMARSAT 3,EMS and ARTEMIS, use advanced on- filters with the special property that adjacent filters, board processing to make efficient use of the available when operated simultaneously, add to form acontinuous L-hand spectrum. In all of these cases, high response without distortion in the crossover (guard performance surface acoustic wave (SAW) devices are band) region. Therefore, when a group of adjacent used. SAW tilters can provide high selectivity (100-200 filters are allocated to a single beam, the entire band kHz transition widths), combined with flat amplitude covered by the filters is usable, without any loss to and linear phase characteristics; their simple intermediate guard bands. construction and radiation hardness also makes them especially suitable for space applications. An overview of SAW based processor architectures is given in section 2.0 of this paper, and the Iladeoffs This paper gives an overview of the architectures used associated with the SAW filters are discussed in Section in the above systems, describing the technologies 3.0. employed, and tile use of bandwidth switchable SAW filtering (BSSF). The tradeoffs to be considered when 2.0 SYSTEM ARCHITECTURES specifying a SAW based system are analyzed, using both theoretical and experimental data. Empirical rules Figure 1 shows a simplified schematic of the for estimating SAW filter performance are given. INMARSAT 3 forward processor, proposed by Matra Achievable performance is illustrated using data from Marconi Space (MMS) and built by COM DEV, while the INMARSAT 3engineering model (EM) processors. Figure 2 shows an exploded view of its physical realization. The return processor is essentially similar, 1.0 INTRODUCTION except for the reversal of the signal paths, and the addition of programmable gain in the individual filter All L-band mobile communication systems must operate channels. within 34 MHz spectrum allocations (1525-1559 MHz forward link, 1626.5-1660.5 MHz return link), and must The key parameters for the INMARSAT 3 processor be able to service low gain mobile terminals. To cope are: with these limitations, systems such as INMARSAT 3, EMS and ARTEMIS use multiple spot beams, frequency Channel bandwidths from 4.5 to 0.45 MHz re-use, and flexible frequency allocation between beams. 20 dB Noise Figure Intermodulation products <-45 dBc These systems require complex on-board processors, 35 dB Nominal gain which use combinations of splitters, amplifiers, SAW 40 dB of programmable gain filters and switch maWices to route traffic to the Maximum mass 35 Kg (total of forward and appropriate beams. Of these processors, which are return processors) currently under development at COM DEV, Maximum power consumption 100 W (total of INMARSAT is by far the most sophisticated, though forward and return processors) ARTEMIS has the most selective filters. The High spectral efficiency (200 kHz guard bands, INMARSAT system also makes limited use of a BSSF) technique called bandwidth switchable SAW filtering 53 Dual redundant right and left circularly polarized chosen as a compromise between minimizing operating (RHCP and LHCP) L-band input signals are split frequency and minimizing fractional bandwidth. After between a total of 15 filter modules, where they are down conversion, the IF signals are amplified by down converted to a 160 MHz IF. Each filter module discrete bipolar amplifiers optimized for low power contains SAW filterbanks to channelize the spectrum, consumption. The signals are then applied to the inputs followed by GaAs FET switch matrices which allow any of the two SAW filterbanks, each of which may contain filter output to be routed to any one of eight output up to three channels. Each filterbank output is them beams. The signals are upconverted to the final L-band amplified by discrete amplifiers and fed into a 3x9 frequency before leaving the filter modules, and are then switch matrix, which allows any channel to be switched combined in the eight output modules (one per beam). to any beam, or to be terminated if not in use. The The mechanical arrangement is forced by the signal switch matrix uses surface mount construction, and is splitting and combining requirements. The input, output built from custom hybridized units each containing three and LO distribution modules are housed in the single pole double throw (SPDT) GaAs FET switches horizontal stack, and interface with the filter modules in and a three way resistive power combiner. Isolation the vertical stack by blind mate connectors; this allows between channels is typically 60 dB. An ASIC controls full connectivity between any input or output module the switch matrix operation. After routing through the and any filter module. Telecommand and telemetry is switch matrix, each of the eight outputs is upconverted handled by the control module, which is placed on top to L-Band. The up conversion frequency is offset from of the vertical (filter) modules; control signals are the down conversion frequency to minimize spurious routed to the horizontal modules via an additional signals. housing on the side of the processor. To minimize mass, all module housings are machined from The LO frequencies are generated externally to the magnesium. processor and are distributed by the LO module. This uses combinations of power splitters and GaAs FET The input modules are among the simplest in the switches to route the LO signals to the appropriate filter system. They contain redundant thin film GaAs input modules. However, the distribution requirements are amplifiers and eight way power dividers implemented extremely complex, and the LO module is with cascaded Wilkinson splitters on high dielectric soft correspondingly complex. substrates. The output modules perform an inverse function, but are considerably more complex. In The EMS system is far simpler in concept than addition to combiners and amplifiers, they contain INMARSAT, though similar technologies are used. It programmable gain blocks implemented with GaAs FET is being built by COM DEV and AME Space for Alenia switches and controlled by an ASIC; interdigital Spazio as a supplementary payload for ITALSAT 2. ban@ass filters are used to remove mixer spurious. The schematic of the EMS forward processor is shown in Figure 4. A Ku band uplink is employed, rather than The f'dter modules, shown schematically inFigure 3, are the C-band uplink used for INMARSAT. Three 4 MHz the key elements in the system, as these provide all the wide slots are selected and down converted to an IF of frequency selectivity and signal routing. Three main approximately 145 MHz, where they are channelized by types of filter module are employed, which differ in the a non-contiguous bank of SAW filters with 250 kHz frequency and bandwidth of their SAW filters, though transition widths. The outputs are then upconverted to a guard bandwidth of 200 kHz is used throughout. the L-band channels 1530-1534 MHz, 1540-1544 MHz Each non-redundant module has a specific LO frequency and 1555-1559 MHz, using different LOs for each filter. that determines its position in the 34 MHz frequency band. Redundant modules can use any LO frequency, The EMS return processor replaces each 4 MHz fdter and can therefore substitute for any module of similar with a bank of four 900 kHz trdters, each with type. The implementation of the INMARSAT independent progranunable gain. Selective use of these frequency plan with only three module types is another subchannels allows coordination with other systems example of the use of BSSF. Because of the contiguous using the same frequency bands. The return f'dters have combining, a given fdter bank can realize several centre frequency separations of 1 MHz and transition channelization schemes, allowing greater standardization widths of 200 kHz; BSSF is not employed. This of module types, and hence greater reliability. frequency plan produces overlap between filters, and hence a reduction in the usable l'dter bandwidth when For reasons discussed in Section 3.0, the SAW Filters adjacent filters are operated simultaneously. The must operate at a relatively low IF; 160 MHz was ARTEMIS system is very similar toEMS, but the return 54 filter transition width is reduced to 100 kHz to avoid where B = transition bandwidth from passband to stopband edge. overlap. No attempt is made to recover these remaining T = impulse response length 100 kHz guard bands using BSSF, but his would be a 201og((l+Sp)/(1-Sp)) = Passband ripple logical extension for future systems. (dB) 3.0 SAW FILTER TECHNOLOGY FOR ON- 201og(8,) = Stopband level (dB) BOARD PROCESSING SAW falters are particularly well suited to the high In the great majority of designs T is 2-3 times the selectivity, linear phase requirements in on-board reciprocal of B. It should also be noted that T is determined by the transition width, and is virtually processing. However their characteristics are very different from those of classical filters, and this often independent of absolute bandwidth; it is also causes confusion when systems are specified. This independent of centre frequency. The physical size of section discusses the tradeoffs and limitations associated the fdter can be obtained by multiplying T by the SAW with this class of SAW f'dter, based on both theoretical velocity. However, the final size is significantly greater than this estimate for two reasons: first, the response and empirical data. must be factored between the two transducers in a non- Reference [1] discusses the basic properties of SAW optimal way, and second, a reasonable separation must be allowed between transducers toavoid electromagnetic filters for mobile communication systems, including BSSF. The SAW transversal filters used in coupling. INMARSAT, EMS and ARTEMIS, all use in-line transducer structures [1]. The transducers contain The choice of factorization is forced by practical considerations. For an in-line transducer structure the numerous interdigitated electrodes (typically 3000 to 9000), formed by photolithography in a thin (1000- allowable weighting pattern on one transducer is restricted so that each electrode covers either all or none 2000A) aluminium film deposited on the polished surface of a piezoelectric crystal; ST-X quartz is used of the aperture (withdrawal weighting). Without this, for these systems on account of its temperature stability. the overall response would not, even to first order, be Each transducer has an ideal frequency response similar the product of the individual transducer responses, and to that of a finite impulse response (FIR) digital filter; this defeats all existing synthesis procedures. the electrodes serve as the taps, and their weights are Empirically, it is well established that individual transducers rarely provide more than 35 dB of close-in controlled by varying the overlaps (apodization). The SAW propagation time between electrodes is equivalent rejection. To achieve higher rejections than this both transducers must contribute significantly to the out of to the sampling time. band response. The withdrawal weighted transducer is The transfer functions of SAW transversal (or FIR) therefore chosen to have reasonable out of band filters have no poles in the finite s plane, only zeros. rejection and a reasonably regular passband response. They are usually also of very high order compared to The apodized transducer can then be designed to satisfy classical f'dters (10000 electrodes in a transducer is not the overall specification. The design is then optimized to correct for second order effects, such as SAW uncommon). Design techniques are therefore quite different, and are usually based on optimization diffraction and circuit loading, but corrections are techniques. Of these, linear programming [1] offers applied to the apodized transducer alone, the other transducer is left fixed; this procedure is most effective unrivalled flexibility. Current programs based on linear if the length of the withdrawal weighted transducer is programming can design both filters and filterbanks with minimized. These design constraints are incompatible arbitrarily specified amplitude and phase responses. The most common requirement is for linear phase, fiat with fully optimal factorization, and some length penalty must be accepted. In addition, the requirements of BSSF amplitude characteristics, both for the individual and the and of correcting for second order effects also produce combined filter responses. a length penalty. For SAW filters, impulse response length is the most For INMARSAT 3 the specified transition bandwidth is appropriate measure of filter complexity. For a linear 200 kHz for all filters. However, a design value of phase design, a simple empirical rule can be used to 170 kHz was used, allowing 10 kHz margin for predict the impulse response length [3]. temperature drift and ± 10 kHz for manufacturing log (Sp_,) = -1.05 - 1.45 BT (1) tolerances. With design passband ripples and stopband 55 levels of 0.2 dB and 50 dB respectively, equation (1) perturbation caused by the metallisation increases in predicts an impulse response length of 13.701as, proportion to frequency. Unfortunately, there is no equivalent to 4.34 cm for quartz (SAW velocity 3159 precise model available for assessing all tradeoffs; m/s). The actual length is approximately 7.1 cm, however, the following empirical formulas give a including 0.9 cm spacing between transducers. The net reasonable estimate of the achievable P-P passband effect of all the above constraints is therefore to ripples for an individual high selectivity quartz filter: increase the total transducer length by about 40% from the estimate given by equation (1). P-P amplitude ripple=Design ripple + 15tF2/B dB (2) A 100 kHz transition width is specified for the P-P phase ripple=Design tipple + 250tF2/B deg (3) ARTEMIS return filters, and a 75 kHz value has been used in the design. Combined with a 0.25 dB passband where t _ metallization thickness (m) (typically ripple and a 50 dB stopband level, equation (I) gives a le-7 to 2e-7 m) predicted impulse duration of 30.3_ (9.56 cm on F= centre frequency (MHz) quartz). The length of the final design is 12.6 cm B= transition bandwidth (MHz) ....... including 1 cm transducer separation. The net transducer length is therefore 21% greater than the ideal The centre frequency should therefore be kept as low as limit. This difference between the ARTEMIS and possible, compatible with the fractional bandwidth INMARSAT filters reflects the absence of BSSF constraints for the material; for filters with transition constraints, and the use of a more sophisticated widths less than 200 kHz, 200 MHz is a reasonable factorization procedure for the ARTEMIS designs. A upper limit. reasonable practical estimate of overall filter length can therefore be obtained by taking the value of T from So far, the effect of shape factor (ratio of bandwidth at equation (1), increasing this by 30%, multiplying by the stopband edges to bandwidth at passband edges) has not SAW velocity, and adding the transducer separation been considered; it does not directly affect device size (0.5-1.0 cm) and an allowance for packaging (0.5-1 cm). but it does have a slight effect on passband ripple and out of band rejection. A low shape factor (very square Manufacturing sensitivity is a critical factor in response) is more difficult to realize with a withdrawal determining the minimum transition bandwidth and weighted transducer, and the overall filter rejection is maximum operating frequency of a SAW filter. reduced as a result. For shape factors of 1.2 or greater, Photolithographic capabilities will allow operation at 1 close in rejections of 50 dB and far out rejections of 60 GHz and above, but the achievable filter performance is dB are achievable. For shape factors of 1.I, these severely degraded, and high selectivity, high precision values are reduced to 45 dB and 50 dB respectively. filters are restricted to comparatively low frequencies. Achievable rejection is also weakly dependent on centre The major limiting factors are: frequency. Metallization uniformity 4.0 EXPERIMENTAL SYSTEM Electrode linewidth control PERFORMANCE Photomask aberrations Substrate uniformity Figure 6 shows the combined response of two L-band Substrate mounting stresses channels measured on the INMARSAT EM forward processor shown in Figure 5. The individual filters All of these produce similar effects, which may be have bandwidths of 0.75 MHz and 2.11 MHz, and modelled as a variation in SAW propagation velocity. together with a 0.54 MHz device form a contiguous set If such velocity errors are random, and average out over of three filters; including the guard band they give a a short distance scale, they are comparatively harmless. total bandwidth of 3.06 MHz. Figure 7 shows the However, the above effects usually produce troublesome response of the 2.11 MHz filter combined with its other long range variations. neighbouring filter to give a net bandwidth of 2.85 MHz. This demonstrates that BSSF can provide For a given effective velocity error, the filter distortion characteristics that are virtually indistinguishable from is directly proportional to centre frequency. If the peak those of individual filters. In the above measurements to peak velocity variations are similar for different filter the unused channels were switched to other outputs lengths, then the distortion is also inversely proportional (beams); the absence of any residual responses to the transition bandwidth. In addition, the velocity 56 demonstrates the low levels of leakage in the SAW [2] O. Andreassen, "A SAW Filter Bank for package, the switch matrix, and the splitter driving the Telecommunications applications", Microwave output mixers. and RF Engineering, pp43-48, Jan/Feb 1990. Figures 8 and 9 show the combined in-band amplitude [3] D.P. Morgan, "Surface-Wave Devices for Signal and phase responses of the three filters. The overall Processing", Amsterdam: Elsevier, 1985. amplitude ripple is approximately 0.5 dB P-P. Mthough not observable in this case, some crossover distortion usually arises, and the ripple in the crossovers is often titvEIl_ i a few tenths of a dB worse than in other regions. The phase ripple clearly shows the transitions between the individual filters. This ripple could be improved by further alignment; but this is not justified as the phase _, :_¢>--qL"I; requirements are comparatively non-critical. The filters are all made in matched sets and little change is observed in passband characteristics over the operating temperature range (-15 to 75°C). tt._* '.-_{>--ql_"I: ihF_rr_ 5.0 CONCLUSIONS The development of the INMARSAT and EMS systems has clearly demonstrated the feasibility of using SAW *1.............. 1.............. based on-board processors for spectrum allocation and routing. It has also provided a great deal of valuable Figure 1. Simplified schematic of information about the tradeoffs associated with the INMARSAT 3 forward )rocessor various technologies, particularly the SAW falters. The greatest technical challenges have not been associated with individual components, but rather with _ the integration of so many technologies into a complete system. Other difficulties have only become fully evident during system level testing. Particularly notable in this regard is the control of spurious signals. The ,,)I- large number of signals and LOs going into the processors, and the large number of leakage paths and non-linear components, make spurious generation a major problem; work is still in progress to isolate and suppress unwanted signals. S ACKNOWLEDGEMENTS The authors would like to acknowledge the invaluable Figure 2. INMARSAT 3 forward processor contributions of P. Kenyon, R. Kovac, B. Van Osch and construction A. Veenstra to this work. tMe, 6.0 REFERENCES _¢*¢r [1] R. Peach & A. Malarky, "Enhanced Efficiency using Bandwidth switchable SAW Filtering for Mobile Satellite Communications Systems", Proc. IMSC, pp394-402, 1990. Figure 3. Simplified schematic of INMARSAT filter module 57 e_A _m TT T Asv 4v cuo Figure 4: Schematic of EMS filter module i ' i . i , i Figure 5. INMARSAT 3 EM forward processor 2OOO0 -- .toI_o t ' _OO_ i i H i ODO0 : ! d ! : -_00¢0 * !....... [....... ._ _ /_ -_o_ _._ _i1i_ I Figure 6. Combined frequency response of channels D and E _1 Figure 7. Combined frequency response of channels E and F .......... -:.- : I :--.: ......:...... :..... g 15(_13 .... ;..... •:...... •,...... ,..... ;..... ;...... :..... 32_ ............. :...... :...... tt..... !...... :...... _...... 10(130 31 50(_ 0.000 _o111 11 -1(1000 i 27000 -15000 ............i.................... .... :...... ....... ;...... :...... i.... ._ -- _ _--q_i Figure 8. Combined amplitude response of channels D, E and F Figure 9. Combined phase restm_e of channels D, E and F 58